Switching power supply apparatus

ABSTRACT

In a switching power supply apparatus, a first switching element is controlled by a driving voltage output from a switching control IC. A second switching control circuit controls the on-time of a second switching element so that the time ratio of the on-time of the second switching element to the on-time of the first switching element becomes almost constant with respect to a change in a load current. In a normal load state, since a square wave output from a frequency setting unit within the switching control IC is output with no change, a converter operates in a current-continuous mode. In a light load state, a driving signal generation unit within the switching control IC is subjected to blanking with the period of a signal output from a maximum frequency setting unit and an oscillation frequency is reduced. Accordingly, the converter operates in a current-discontinuous mode.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a switching power supply apparatus thatmaintains high power conversion efficiency under a condition ofextensive power supply to a load.

2. Description of the Related Art

In the past, in order to reduce electric power consumption during astandby time of a load circuit serving as a supply destination of apower-supply voltage, i.e., standby power consumption, there has been aswitching power supply apparatus causing a converter for a normal timeand a converter for a standby time to operate by switching therebetweenor a switching power supply apparatus switching between operation modesat a normal time and a standby time in a single converter.

The following three types of switching power supply apparatuses havebeen used as the switching power supply apparatus switching between theoperation modes in the above-mentioned single converter.

The first type is a switching power supply apparatus where, in a flybackconverter or the like, at a standby time (i.e., at a time of a lightload), the constant of a circuit is changed, the switching frequency ofa switching element is reduced, and the number of switching operationsper unit time is reduced, thereby reducing a switching loss.

The second type is a switching power supply apparatus where, in acurrent resonant converter or the like, at a standby time, due to anintermittent oscillation operation in which an oscillation time periodand a halt time period are repeated with an oscillation frequency at ahigh frequency operation being maintained, the number of switchingoperations per unit time is reduced and a switching loss is reduced,thereby achieving high efficiency.

The third type is a switching power supply apparatus where, at a standbytime, by decreasing an output voltage, the power-supply voltage of aswitching control circuit is reduced, thereby reducing electric powerconsumption occurring in the switching control circuit or the like.

FIG. 1 is the circuit diagram of a switching power supply apparatus ofthe third type and disclosed in Japanese Unexamined Patent ApplicationPublication No. 7-274498. In FIG. 1, when the light load state of a load8 has been continued for a predetermined time, the operation mode of acontrol circuit 9 is switched to an electric power saving mode by anoperation mode switching mechanism 50, the control frequency of afrequency control mechanism 23 is controlled so as to be high, and thepower-supply voltage Vcc of the control circuit 9 is decreased to benear to a minimum allowable operating voltage. Accordingly, in the lightload state, the control circuit 9 operates in the vicinity of theminimum allowable operating voltage, thereby reducing electric powerconsumption, and the efficiency of a resonance-type switching powersupply apparatus is improved.

In the first type, usually a converter is configured using one switchingelement, a large switching surge voltage occurs at both ends of theswitching element, in a rated load, and it is necessary for theswitching element to have a high-voltage characteristic. In general,since the cost of a high-voltage switching element is high and a voltagedrop at the time of conduction is large, a conduction loss is large. Asa result, there exists a problem that efficiency in the rated load islowered.

In the second type, since the intermittent oscillation operation isadopted, there exists a problem that the ripple of an output voltage islarge or the period of intermittent oscillation existing in an audiblefrequency range results in making a sound. Furthermore, since, in anoscillation time period, two FETs operate, there exists a problem thatthe drive loss of the FET is large or an output response characteristicwith respect to the rapid fluctuation of an output load is poor.

In the third type, since a switching frequency becomes high while thepower consumption of the switching control circuit or the like isreduced at the standby time, a switching loss becomes large, and theeffect of reducing power consumption is small.

SUMMARY OF THE INVENTION

Preferred embodiments of the present invention provide a switching powersupply apparatus that achieves a high-efficiency operation with respectto the wide fluctuation of a load ranging from a light load, to a ratedload, and to a heavy load and is superior in an output responsecharacteristic.

A switching power supply apparatus according to a preferred embodimentof the present invention includes a direct-current power-supply inputunit to which a direct-current input voltage is input, a transformerconfigured to include at least a primary winding and a secondary windingmagnetically coupled to each other, a first series circuit connected toboth ends of the direct-current power-supply input unit and in which theprimary winding, an inductor, and a first switching element areconnected in series, a second series circuit connected to both ends ofthe first switching element or both ends of a series circuit includingthe primary winding and the inductor and in which a second switchingelement and a capacitor are connected in series, a switching controlcircuit configured to put the second switching element into an on-stateduring a predetermined time after having put the first switching elementinto an on-state during a predetermined time, and a first rectificationsmoothing circuit configured to rectify and smooth analternating-current voltage output from the secondary winding and outputan output voltage to a load, wherein the switching power supplyapparatus includes a light load detection mechanism configured todirectly or indirectly detect a load current flowing through the loadand to detect whether or not the load is in a light load state, theswitching control circuit includes a first switching control circuitconfigured to control an on-time of the first switching element, and asecond switching control circuit configured to control an on-time of thesecond switching element so that an on-time ratio serving as a ratio ofthe on-time of the second switching element to the on-time of the firstswitching element becomes almost constant with respect to a change inthe load current, in a normal load state, an operation is performed in acurrent-continuous mode where the first switching element and the secondswitching element are complementarily and repeatedly put into on-statesand off-states, and when the light load detection mechanism has detectedthe light load state, an operation is performed in acurrent-discontinuous mode where a current halt time period in which nocurrent flows through the transformer exists between turnoff of thesecond switching element and turnon of the first switching element.

In addition, a switching power supply apparatus of a preferredembodiment of the present invention preferably includes a direct-currentpower-supply input unit to which a direct-current input voltage isinput, a transformer configured to include at least a primary windingand a secondary winding, magnetically coupled to each other, a firstseries circuit in which a first switching element and a second switchingelement, connected to both ends of the direct-current power-supply inputunit, are connected in series, a second series circuit configured to beconnected to both ends of the second switching element and in which theprimary winding, an inductor, and a capacitor are connected in series, aswitching control circuit configured to put the second switching elementinto an on-state during a predetermined time after having put the firstswitching element into an on-state during a predetermined time, and afirst rectification smoothing circuit configured to rectify and smoothan alternating-current voltage output from the secondary winding andoutput an output voltage to a load, wherein the switching power supplyapparatus includes a light load detection mechanism configured todirectly or indirectly detect a load current flowing through the loadand detect whether or not the load is in a light load state, theswitching control circuit includes a first switching control circuitconfigured to control an on-time of the first switching element, and asecond switching control circuit configured to control an on-time of thesecond switching element so that an on-time ratio serving as a ratio ofthe on-time of the second switching element to the on-time of the firstswitching element becomes almost constant with respect to a change inthe load current, in a normal load state, an operation is performed in acurrent-continuous mode where the first switching element and the secondswitching element are complementarily and repeatedly put into on-statesand off-states, and when the light load detection mechanism has detectedthe light load state, an operation is performed in acurrent-discontinuous mode where a current halt time period in which nocurrent flows through the transformer exists between turnoff of thesecond switching element and turnon of the first switching element.

The transformer preferably includes a first drive winding, and a secondrectification smoothing circuit that supplies a direct-currentpower-supply voltage to the switching control circuit is provided in thefirst drive winding.

A timing detection mechanism configured to detect a timing when thesecond switching element is turned off and a winding voltage of thetransformer is inverted and a feedback circuit configured to generate afeedback signal used to detect and control an output voltage areincluded, wherein the first switching control circuit is a circuit thatcauses the first switching element to be turned on with a slight delayafter the timing detected by the timing detection mechanism and causesthe first switching element to be turned off after a time (the on-timeof the first switching element) corresponding to a voltage of thefeedback signal has elapsed.

For example, the timing detection mechanism preferably detects a voltagechange occurring between both ends of the winding voltage of thetransformer.

The timing detection mechanism preferably detects a change in adrain-to-source voltage of at least one of the first switching elementand the second switching element.

The timing detection mechanism may also preferably detect a change in adrain current or a source current of at least one of the first switchingelement and the second switching element.

The timing detection mechanism may further preferably detect a voltagechange occurring between both ends of the first drive winding.

The transformer includes a second drive winding, a connection point ofthe first switching element with the second switching element and afirst end of the second drive winding are connected to each other, thesecond switching control circuit is connected between a second end ofthe second drive winding and a control terminal of the second switchingelement, and the second switching control circuit is a circuit thatcauses the second switching element to be turned on, using a voltage ofthe second drive winding, which occurs due to turnoff of the firstswitching element.

The second switching control circuit preferably is a circuit thatperforms control so that the on-time ratio becomes almost one.

When the on-time ratio, the direct-current input voltage, the outputvoltage, the number of turns of the primary winding, and the number ofturns of the secondary winding are expressed with Da, Vi, Vo, np, andns, respectively, the second switching control circuit is preferably acircuit that causes the second switching element to be turned off byperforming control so that the Da becomes larger than ns·Vi/np·Vo.

The second switching control circuit is a circuit that preferably causesthe second switching element to be turned off by performing control sothat the Da becomes larger than ns·Vi/np Vo−1.

The light load detection mechanism preferably performs detection using avoltage of the feedback signal.

The light load detection mechanism also preferably performs detection onthe basis of whether or not a switching frequency to drive the firstswitching element or the second switching element has reached an upperlimit value.

In the current-discontinuous mode, along with a decrease in the loadcurrent, the switching control circuit preferably causes a switchingfrequency to drive the first switching element or the second switchingelement to decrease.

In the current-discontinuous mode, along with a decrease in the loadcurrent, the switching control circuit preferably reduces the on-time ofthe first switching element or the second switching element.

In the current-continuous mode, the switching control circuit preferablysets a maximum level of the switching frequency or a maximum value ofthe on-time of the first switching element or the second switchingelement.

In the current-discontinuous mode, the switching control circuitpreferably controls the switching frequency or the on-time of the firstswitching element or the second switching element, in response to thefeedback signal.

The switching control circuit preferably sets a limit value on a voltageof the feedback signal, and causes switching of the first switchingelement or the second switching element to be halted when the voltage ofthe feedback signal has exceeded the limit value.

The switching control circuit preferably operates with the lowestswitching frequency and a minimum on-time, with respect to the limitvalue.

In the current-discontinuous mode, while the switching control circuitdoes not cause the second switching element to be turned on even if thefirst switching element has been turned off, the switching controlcircuit preferably applies a forward current to a parasitic diode of thesecond switching element first, and applies a backward current during asubsequent reverse recovery time.

At least one rectifying element provided in the first switching element,the second switching element, or the first rectification smoothingcircuit preferably is a field-effect transistor.

The inductor preferably includes a leakage inductance of thetransformer.

According to various preferred embodiments of the present invention, thefollowing advantageous effects are obtained.

In the light load state such as the standby time, an operation isperformed in the standby mode where the switching frequency is reducedand an operation is performed in the current-discontinuous mode.Accordingly, the number of switching operations per unit time isreduced, the switching loss is significantly reduced and prevent, andhence, it is possible to achieve high efficiency.

Since, in the light load state, the second switching element (high sideFET) is put into an on-state for enough time to absorb a surge voltageoccurring in the first switching element Q1 when the first switchingelement (low side FET) is turned off, it is possible to reduce awithstand voltage required for the first switching element Q1.

Since an intermittent oscillation operation is avoided, there is noincrease in an output ripple, and the voltage accuracy of the outputvoltage is excellent.

Since an oscillation halt time period in such a way as intermittentoscillation does not exist, a response characteristic with respect tothe rapid change of a load is excellent.

Since, in the light load state such as the standby time, an electricpower conversion operation is performed due to almost only the operationof the first switching element (low side FET), it is possible to reducethe drive loss of an FET.

The above and other elements, features, steps, characteristics andadvantages of the present invention will become more apparent from thefollowing detailed description of the preferred embodiments withreference to the attached drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a circuit diagram of a switching power supply apparatusdisclosed in Japanese Unexamined Patent Application Publication No.7-274498.

FIG. 2 is a circuit diagram of a switching power supply apparatus 101according to a first preferred embodiment of the present invention.

FIG. 3 is a circuit diagram of a feedback circuit 83 in FIG. 2.

FIG. 4 is a block diagram illustrating a configuration between an FBterminal and an OUT terminal within a switching control IC 81.

FIG. 5 is a diagram illustrating a change in a switching frequency dueto a change in a weight of a load, in a normal load state.

FIG. 6 is a diagram illustrating a change in a switching frequency and achange of a mode when a transition from a normal load state to a lightload state has occurred.

FIG. 7 is a diagram illustrating a relationship between a voltage of theFB terminal and a blanking frequency within the switching control IC 81.

FIG. 8 is a circuit diagram of a switching power supply apparatus 102according to a second preferred embodiment of the present invention.

FIG. 9 is a circuit diagram of a switching power supply apparatus 103according to a third preferred embodiment of the present invention.

FIG. 10 is a waveform diagram illustrating a relationship among agate-to-source voltage Vgs1 of a first switching element Q1, agate-to-source voltage Vgs2 of a second switching element Q2, adrain-to-source voltage Vds1 of the first switching element Q1, and avoltage of a capacitor Cb2.

FIG. 11 is a circuit diagram of a switching power supply apparatus 104according to a fourth preferred embodiment of the present invention.

FIG. 12 is a waveform diagram illustrating a relationship among agate-to-source voltage Vgs1 of a first switching element Q1, agate-to-source voltage Vgs2 of a second switching element Q2, adrain-to-source voltage Vds1 of the first switching element Q1, and avoltage Vcb2 of a capacitor Cb2 in FIG. 11.

FIG. 13 is a circuit diagram of a switching power supply apparatus 105according to a fifth preferred embodiment of the present invention.

FIG. 14 is a waveform diagram illustrating a relationship among agate-to-source voltage Vgs1 of a first switching element Q1, agate-to-source voltage Vgs2 of a second switching element Q2, adrain-to-source voltage Vds1 of the first switching element Q1, and avoltage Vcb2 of a capacitor Cb2 in FIG. 13.

FIG. 15 is a circuit diagram of a switching power supply apparatus 106according to a sixth preferred embodiment of the present invention.

FIG. 16 is a waveform diagram illustrating a relationship among agate-to-source voltage Vgs1 of a first switching element Q1, agate-to-source voltage Vgs2 of a second switching element Q2, adrain-to-source voltage Vds1 of the first switching element Q1, and avoltage Vcb2 of a capacitor Cb2 in FIG. 15.

FIG. 17 is a circuit diagram of a switching power supply apparatus 107according to a seventh preferred embodiment of the present invention.

FIG. 18 is a circuit diagram of a switching power supply apparatus 108according to an eighth preferred embodiment of the present invention.

FIG. 19 is a circuit diagram of a switching power supply apparatus 109according to a ninth preferred embodiment of the present invention.

FIG. 20 is a circuit diagram of a switching power supply apparatus 110according to a tenth preferred embodiment of the present invention.

FIG. 21 is a circuit diagram of a switching power supply apparatus 111according to an eleventh preferred embodiment of the present invention.

FIG. 22 is a circuit diagram of a switching power supply apparatus 112according to a twelfth preferred embodiment of the present invention.

FIG. 23 is a circuit diagram of a switching power supply apparatus 113according to a thirteenth preferred embodiment of the present invention.

FIG. 24 is a circuit diagram of a switching power supply apparatus 114according to a fourteenth preferred embodiment of the present invention.

FIG. 25 is a circuit diagram of a switching power supply apparatus 115according to a fifteenth preferred embodiment of the present invention.

FIG. 26 is a circuit diagram of a switching power supply apparatus 116according to a sixteenth preferred embodiment of the present invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS First PreferredEmbodiment

A switching power supply apparatus according to a first preferredembodiment will be described with reference to FIG. 2 to FIG. 7.

FIG. 2 is the circuit diagram of a switching power supply apparatus 101according to the first preferred embodiment. The voltage of adirect-current input power supply Vi is input between the inputterminals PI(+) and PI(G) of this switching power supply apparatus 101.In addition, a predetermined direct-current voltage is output to a loadRo connected between the output terminals PO(+) and PO(G) of theswitching power supply apparatus 101.

Between the input terminals PI(+) and PI(G), a first series circuit isprovided and includes a first switching element Q1, a second switchingelement Q2, and a current detection resistor Ri connected in series. Thefirst switching element Q1 and the second switching element Q2 includeFETs, the drain terminal of the high-side second switching element Q2 isconnected to the input terminal PI(+), and the source terminal of thelow-side first switching element Q1 is connected to the input terminalPI(G) through the current detection resistor Ri.

In both ends of the second switching element Q2, a second series circuitis provided and includes a capacitor Cr, the primary winding np of atransformer T, and an inductor Lr connected in series. The inductor Lris configured using the leakage inductance of the primary winding np ofthe transformer T.

In the secondary windings ns1 and ns2 of the transformer T, a firstrectification smoothing circuit is provided and includes diodes Ds andDf and a capacitor Co. This first rectification smoothing circuitfull-wave rectifies, smooths, and outputs an alternating-currentvoltage, output from the secondary windings ns1 and ns2, to the outputterminals PO(+) and PO(G).

A rectification smoothing circuit based on a diode Db and a capacitor Cbis connected to the first drive winding nb1 of the transformer T. Adirect-current voltage obtained from the rectification smoothing circuitis supplied between the GND terminal and the VCC terminal of a switchingcontrol IC 81, as a power-supply voltage.

The switching control IC 81 corresponds to a first switching controlcircuit according to a preferred embodiment of the present invention.The switching control IC 81 outputs a driving voltage Vgs1 from the OUTterminal thereof to the gate of the first switching element Q1. Thefirst switching element Q1 performs an on-off operation due to theabove-mentioned driving voltage Vgs1.

A second switching control circuit 61 serving as a high side switchingcontrol circuit is provided between the second drive winding nb2 of thetransformer T and the second switching element Q2. Specifically, thefirst end of the second drive winding nb2 of the transformer T isconnected to a connection point between the first switching element Q1and the second switching element Q2 (the source terminal of the secondswitching element Q2), and the second switching control circuit 61 isconnected between the second end of the second drive winding nb2 and thegate terminal of the second switching element Q2.

Using the voltage of the second drive winding nb2, which occurs due tothe turnoff of the first switching element Q1, the second switchingcontrol circuit 61 causes the second switching element Q2 to be turnedon. After the second switching element Q2 has been turned on, theon-time of the second switching element Q2 is controlled so that theratio of the on-time (ton2) of the second switching element Q2 to theon-time (ton1) of the first switching element Q1, namely, an on-timeratio Da (Da=ton2/ton1) becomes almost constant with respect to a changein a load. For example, if the Da is approximately one, the on-times ofthe Q1 and the Q2 are approximately equal to each other.

A feedback circuit 83 is provided among the output terminals PO(+) andPO(G) and the switching control IC 81. This feedback circuit 83 is acircuit that generates a feedback signal due to the comparison of thevoltage-dividing value of a voltage between the output terminals PO(+)and PO(G) with a reference voltage and inputs a feedback voltage to theFB terminal of the switching control IC 81, in an insulating state.

FIG. 3 is the circuit diagram of the feedback circuit 83. Between theoutput terminals PO(+) and PO(G), a series circuit based on a shuntregulator SR, a resistor R3, and a light emitting element of a photocoupler PC and a voltage-dividing circuit based on resistors R1 and R2are connected. The voltage-dividing output of the above-mentionedvoltage-dividing circuit based on the resistors R1 and R2 is provided tothe reference terminal of the shunt regulator SR. In addition, betweenthe voltage control end and the reference terminal of the shuntregulator SR, a negative feedback circuit including a resistor R11 and acapacitor C11 is provided. In addition, the light receiving element ofthe photo coupler PC is connected between the FB terminal and the GNDterminal of the switching control IC 81. Within the switching control IC81, a constant current circuit is connected to the FB terminal.

The feedback circuit 83 functions according to a relationship where whenan output voltage to the output terminals PO(+) and PO(G) becomes higherthan a setting voltage, the voltage of the FB terminal decreases withthat extent.

FIG. 4 is a block diagram illustrating a configuration between the FBterminal and the OUT terminal within the switching control IC 81. Inresponse to the feedback voltage input to the FB terminal, a frequencysetting unit 91 sets the frequency (period) of a driving signalgenerated by the driving signal generation unit 92. The driving signalgeneration unit 92 outputs the driving voltage Vgs1 from the OUTterminal to the gate of the first switching element Q1.

The frequency setting unit 91 generates a square-wave signal in arelationship where a switching frequency increases with a decrease inthe voltage of the FB terminal, and outputs the square-wave signal tothe driving signal generation unit 92.

In a load state that is not a light load state such as a standby timeand in which an operation is performed in a normal operation mode, whena load is reduced and the output voltage becomes higher than the settingvoltage, for example, the switching frequency increases with thatextent. Thus, the on-time of the first switching element Q1 isshortened.

So as to prevent the switching frequency from becoming too high, amaximum frequency setting unit 93 sets the upper limit value (blankingfrequency) of the switching frequency. The driving signal generationunit 92 is subjected to blanking with the period of a signal output fromthe maximum frequency setting unit 93. In a normal load state in whichan operation is performed in the normal operation mode, since a squarewave output from the frequency setting unit 91 is output with no change,a converter operates in a current-continuous mode. On the other hand, inthe light load state in which an operation is performed in a standbyoperation mode, the period of a signal output from the driving signalgeneration unit 92 is controlled on the basis of the signal output fromthe maximum frequency setting unit 93, so as not to become shorter thana setting value. Therefore, in a state where this restriction isimposed, while the on-times of the first switching element Q1 and thesecond switching element Q2 become short, a blank time is insertedbetween a time when the Q2 is turned off and a time when the Q1 isturned on next. Accordingly, the converter operates in acurrent-discontinuous mode.

FIG. 5 is a diagram illustrating a change in a switching frequency dueto a change in the weight of a load, in the load state of a normaloperation mode.

As described above, after the second switching element Q2 has beenturned on, the second switching control circuit 61 in FIG. 2 controlsthe on-time of the second switching element Q2 so that the on-time ratioDa becomes almost constant. FIG. 5 corresponds to a case where the Da isapproximately one. In this regard, however, a dead time is providedbetween the on-time period of the first switching element Q1 and theon-time period of the second switching element Q2.

In this way, in the normal load state, an operation is performed in thecurrent-continuous mode, in a relationship where the switching frequencyincreases with a decrease in the load.

FIG. 6 is a diagram illustrating a change in the switching frequency andthe change of a mode when a transition from the normal load state to thelight load state has occurred.

In the normal load state, an operation is performed in thecurrent-continuous mode in such a way as described above, and theswitching frequency increases with a decrease in the load. After theswitching frequency has reached the blanking frequency, even if the loadbecomes lighter, the frequency of the driving voltage output from thedriving signal generation unit 92 illustrated in FIG. 4 does not becomegreater than or equal to the maximum frequency, due to the effect of themaximum frequency setting unit 93. Therefore, in the light load statewhere the blanking frequency is reached, while the on-times of theswitching elements Q1 and Q2 decrease with a decrease in the load, ablank time is inserted during a period up to the turn-on of theswitching elements Q1 and Q2. As a result, in the light load state, anoperation is performed in the current-discontinuous mode.

FIG. 7 is a diagram illustrating a relationship between the voltage ofthe above-mentioned FB terminal and a blanking frequency. In thisexample, when the voltage of the FB terminal is lower than about 0.4 V,for example, (a no-load state or an approximate no-load state), aswitching operation is halted. In an interval in which the voltage ofthe FB terminal is about 0.4 V to about 0.6 V, for example, an operationis performed with the switching frequency of 1 kHz, and in an intervalin which the voltage of the above-mentioned FB terminal is about 0.6 Vto about 1.0 V, for example, the switching frequency linearly changes.When the voltage of the FB terminal is higher than about 1.0 V, forexample, (the normal load state), a maximum blanking frequency becomesabout 250 kHz, for example.

Accordingly, since, in the light load state, the switching frequencydoes not exceed about 250 kHz and the switching frequency furtherdecreases with a decrease in the load, a power conversion loss is evenmore significantly reduced and prevented.

In addition, the blanking frequency may be fixed (for example, about 130kHz) independently of the voltage of the FB terminal. In addition, themaximum on-time of the switching element may be set independently of thevoltage of the FB terminal.

As illustrated above, since, in the light load state, the secondswitching element (high side FET) Q2 is put into an on-state for enoughtime period to absorb the surge voltage occurring when the firstswitching element Q1 (low side FET) is turned off, it is possible toreduce the withstand voltage of the FET.

In addition, since an intermittent oscillation operation is avoided,there is no increase in an output ripple, and the voltage accuracy ofthe output voltage is excellent. Furthermore, since an oscillation halttime period in such a way as intermittent oscillation does not exist, aresponse characteristic with respect to the rapid change of a load isexcellent.

In addition, since, in the light load state such as the standby time,the electric power conversion operation is performed due to almost onlythe operation of the first switching element Q1 (low side FET), it ispossible to reduce a drive loss in the switching element.

In addition, when, in the light load state, a voltage occurring in thesecond drive winding nb2 does not reach a voltage required to turn onthe second switching element Q2, a reverse current flows through aparasitic diode (not illustrated) of the second switching element Q2during the reverse recovery time of the parasitic diode. Accordingly,even if the second switching element Q2 is not put into an on-state, acurrent flows through the parasitic diode of the second switchingelement Q2, first in a forward direction and then in a reversedirection. Accordingly, it is possible to absorb the surge voltageoccurring in the first switching element Q1.

Second Preferred Embodiment

FIG. 8 is the circuit diagram of a switching power supply apparatus 102according to a second preferred embodiment.

While being different from the switching power supply apparatus 101according to the first preferred embodiment illustrated in FIG. 2, FIG.8 specifically illustrates the configuration of a switching control IC84 and the configuration of a second switching control circuit 63.

A series circuit including a constant current circuit CC1 and acapacitor C3 is connected to the OUT terminal of the switching controlIC 84, and connected so that the charging voltage of the capacitor C3 isinput to an IS terminal.

The voltage of a counter electromotive force induced in the first drivewinding nb1 due to the turnoff of the second switching element Q2 isinput to a ZT terminal, and hence, the switching control IC 84 puts theOUT terminal into a high level. Accordingly, the first switching elementQ1 is turned on.

On the basis of the input voltage of the ZT terminal, the switchingcontrol IC 84 detects a timing when the second switching element Q2 isturned off and the winding voltage of the transformer T is inverted.This configuration corresponds to a “timing detection mechanism”according to a preferred embodiment of the present invention.

Due to the voltage of the OUT terminal of the switching control IC 84,the constant current circuit CC1 constant-current charges the capacitorC3. A comparator within the switching control IC 84 compares the voltageof the capacitor C3 with the voltage of the FB terminal, and when thevoltage of the capacitor C3 has become equal to the voltage of the FBterminal, the first switching element Q1 is turned off. Accordingly, thevoltage of the FB terminal decreases with an increase in an outputvoltage to the output terminals PO(+) and PO(G), and hence, the chargingtime of the capacitor C3 is shortened. Accordingly, the on-time of thefirst switching element Q1 is shortened. Due to such a mechanism, theoutput voltage to the output terminals PO(+) and PO(G) is controlled soas to become a constant voltage.

On the other hand, when the first switching element Q1 has been turnedon, the capacitor Cb2 is discharged with a constant current through thepath of the capacitor Cb2→a diode D3→the constant current circuit CC2→adiode D2→the second drive winding nb2 due to a voltage induced in thesecond drive winding nb2, in the second switching control circuit 63,and a negative voltage occurs.

After that, when the first switching element Q1 has been turned off, apositive voltage is applied to the second switching element Q2 through aresistor R4 due to the voltage induced in the second drive winding nb2,and the Q2 is turned on. In addition, the electric charge of thecapacitor Cb2 is constant-current charged in a positive directionthrough the path of the second drive winding nb2→4 a diode D1→theconstant current circuit CC2→a diode D4→the capacitor Cb2. When thevoltage of the capacitor Cb2 has exceeded about 0.6 V serving as thethreshold voltage of a transistor, for example, the transistor Q3 is putinto an on-state, and accordingly, the second switching element Q2 isturned off. The charging time of the capacitor Cb2, i.e., the on-time ofthe first switching element Q1, and the discharging time of thecapacitor Cb2, i.e., the on-time of the second switching element Q2,become equal to each other, and the on-time ratio Da is controlled so asto be approximately one.

Third Preferred Embodiment

FIG. 9 is the circuit diagram of a switching power supply apparatus 103according to a third preferred embodiment. A second switching controlcircuit 62 in the switching power supply apparatus 103 is different fromFIG. 8. In addition, a circuit on a secondary side is different from thecircuits illustrated in FIG. 2 and FIG. 8.

In the second switching control circuit 62, a constant current circuitpreferably includes transistors Q4 and Q5 and resistors R7 and R8.Accordingly, the basic circuit operation of the second switching controlcircuit 62 is the same as the second switching control circuit 63illustrated in FIG. 8. In addition, a series circuit including a diodeD5 and a resistor R6 is connected in parallel to a resistor R4.Therefore, the rise of the driving voltage Vgs2 of the second switchingelement Q2 is set due to the parallel impedance of the R4 and the R6,and the fall thereof is dominantly set only due to the impedance of theR4.

An FET Qs and an FET Qf used for rectification are connected to thesecondary windings ns1 and ns2 of the transformer T. A secondary sideswitching control circuit 86 performs switching of the FET Qs and FET Qfin synchronization with a change in an electromotive voltage occurringin the secondary winding ns1 or ns2 of the transformer T.

In this way, synchronous rectification is performed using a rectifyingFET in place of a rectifier diode, and hence, it is possible to reduce aconduction loss occurring due to the flow of a current.

FIG. 10 is a waveform diagram illustrating a relationship among thegate-to-source voltage Vgs1 of the first switching element Q1, thegate-to-source voltage Vgs2 of the second switching element Q2, thedrain-to-source voltage Vds1 of the first switching element Q1, and thevoltage of the capacitor Cb2 in FIG. 8 or FIG. 9.

When the first switching element Q1 has been put into an on-state, anegative voltage is induced in the high side drive winding nb2, and thecharging voltage VCb2 of the capacitor Cb2 decreases from about 0.6 Vserving as the threshold voltage, for example. After that, when thefirst switching element Q1 has been turned off, a positive voltage isinduced in the high side drive winding nb2, and the charging voltageVCb2 of the capacitor Cb2 increases. When the charging voltage VCb2 ofthis capacitor Cb2 has exceeded about 0.6 V serving as the thresholdvoltage, for example, the transistor Q3 is put into an on-state.Accordingly, the gate electric potential of the second switching elementQ2 becomes 0 V, and the second switching element Q2 is turned off. Sincethe capacitor Cb2 is charged and discharged with the constant current ofa same current value, the slope of the charging voltage VCb2 is equal.Thus, a charge-discharge current ratio Di is 1:1. Therefore, the on-timeof the second switching element Q2 is equal to the on-time of the firstswitching element Q1.

In FIG. 10, T_(Q1ON)(1) and T_(Q2ON)(1) are equal to each other due tothe above-mentioned operation. Here, when the on-time of the firstswitching element Q1 has been lengthened and has become T_(Q1ON)(2), theVds1 and the Vcb2 correspond to a waveform diagram illustrated by dottedlines. Also at this time, the T_(Q1ON)(2) and T_(Q2ON)(2) are equal toeach other due to the above-mentioned operation.

Fourth Preferred Embodiment

FIG. 11 is the circuit diagram of a switching power supply apparatus 104according to a fourth preferred embodiment.

While being different from the switching power supply apparatusillustrated in the first preferred embodiment, a feedback circuit issimply illustrated and only the path of feedback is expressed using oneline (Feed back). In addition, the configuration of a second switchingcontrol circuit 66 is different. In this example, a circuit is providedin which two zener diodes Dz1 and Dz2 are connected in series in areverse direction and furthermore a resistor R7 is connected in series.This circuit inputs thereto the output voltage of the high side drivewinding nb2 and bi-directionally generates a constant voltage. A voltagebetween both ends of the zener diodes Dz1 and Dz2 connected in series isapplied to a charge-discharge circuit based on a capacitor Cb2 and aresistor R8.

In order to cause the currents of the charge and discharge of thecapacitor Cb2 to be equal to each other, the zener voltages of the twozener diodes Dz1 and Dz2 are basically equal to each other.

FIG. 12 is a waveform diagram illustrating a relationship among thegate-to-source voltage Vgs1 of the first switching element Q1, thegate-to-source voltage Vgs2 of the second switching element Q2, thedrain-to-source voltage Vds1 of the first switching element Q1, and thevoltage Vcb2 of the capacitor Cb2 in FIG. 11.

First, when the first switching element Q1 has been turned on, anegative voltage induced in the high side drive winding nb2 isstabilized to be a constant voltage by a series circuit including thezener diodes Dz1 and Dz2 and the resistor R7, and applied to a timeconstant circuit based on the capacitor Cb2 and the resistor R8.

After that, when the first switching element Q1 has been turned off, apositive voltage induced in the high side drive winding nb2 is appliedthrough a resistor R5 and the positive voltage is applied to the secondswitching element Q2. Therefore, the Q2 is turned on. In addition, whenthe positive voltage induced in the high side drive winding nb2 isstabilized to be a constant voltage by the series circuit including thezener diodes Dz1 and Dz2 and the resistor R7, and the above-mentionedstabilized positive voltage is applied to the time constant circuitbased on the capacitor Cb2 and the resistor R8.

A transistor Q3 is put into an on-state at a time when the voltage ofthe capacitor Cb2 exceeds about 0.6 V serving as the threshold voltage,for example, and accordingly the second switching element Q2 is turnedoff.

According to the above-mentioned operation, the charging time of thecapacitor Cb2, i.e., the on-time of the first switching element Q1, andthe discharging time of the capacitor Cb2, i.e., the on-time of thesecond switching element Q2, become approximately equal to each other.

In FIG. 12, T_(Q1ON)(1) and T_(Q2ON)(1) are equal to each other due tothe above-mentioned operation. Here, when the on-time of the firstswitching element Q1 has been lengthened and has become T_(Q1ON)(2), theVds1 and the Vcb2 correspond to a waveform diagram illustrated by dottedlines. Also at this time, the T_(Q1ON)(2) and T_(Q2ON)(2) are equal toeach other due to the above-mentioned operation.

Fifth Preferred Embodiment

FIG. 13 is the circuit diagram of a switching power supply apparatus 105according to a fifth preferred embodiment.

While, in the second to fourth preferred embodiments, the on-time ratioDa (=T_(Q2ON)/T_(Q1ON))=1 is preferably satisfied, the fifth preferredembodiment is an example corresponding to a case where the on-time ratioDa (=T_(Q2ON)/T_(Q1ON))≠1 is preferably satisfied.

While being different from the switching power supply apparatusillustrated in the first preferred embodiment, a circuit configurationon a secondary side is the circuit configuration of a flyback method,and the switching power supply apparatus is configured using arectification smoothing circuit including a diode Ds and a capacitor Co.The configuration of a second switching control circuit 67 is alsodifferent. In this example, two constant current circuits CC21 and CC22,diodes D1 and D2 to prevent backflow, and a capacitor C5 are provided.

When the first switching element Q1 has been turned on, the capacitorCb2 is discharged, due to a voltage induced in the high side drivewinding nb2, in a negative direction with a constant current through thepath of the capacitor Cb2→the constant current circuit CC22→the diodeD2→the high side drive winding nb2.

After that, when the first switching element Q1 has been turned off, apositive voltage is applied to the second switching element Q2 through aresistor R5 and the capacitor C5, due to a positive voltage induced inthe high side drive winding nb2, and the Q2 is turned on. In addition,the capacitor Cb2 is charged in a positive direction with a constantcurrent through the path of the high side drive winding nb2→the diodeD1→the constant current circuit CC21→the capacitor Cb2. The transistorQ3 is put into an on-state at a time when the voltage of the capacitorCb2 exceeds about 0.6 V serving as the threshold voltage, for example,and accordingly the second switching element Q2 is turned off.

Since a charge current and a discharge current to the capacitor Cb2 areindependently defined, the discharging time of the capacitor Cb2, i.e.,the on-time of the first switching element Q1, and the charging time ofthe capacitor Cb2, i.e., the on-time of the second switching element Q2,are different from each other. In this regard, however, a ratio betweenthe charging time and the discharging time for the capacitor Cb2 isconstant, and has a proportional relationship (linear) with the on-timeof the first switching element Q1. Therefore, the on-time ratio Da(=T_(Q2ON)/T_(Q1ON)) is maintained constant due to a preliminarilydefined relationship.

Here, for example, it is assumed that an electric power convertercircuit has a circuit configuration to which a flyback converter isapplied. Furthermore, a case will be described where, after theswitching element Q2 has been turned off, the reset time of atransformer is over, the switching element Q1 is turned on by utilizingthe inversion of a transformer voltage due to the finish of the reset ofthe transformer, and hence, a current flowing through the transformer iscaused to operate in a current critical mode.

If an operation is performed in the current critical mode, it is easy toachieve a zero voltage switching operation. In addition to this, it ispossible to reduce a loss associated with the reverse recovery time of arectifier diode on a secondary side, and it is possible to realize ahighly efficient operation under a specific design condition. However,if an operation is performed in the current critical mode, the on-timeof the switching element Q1 changes greatly with respect to thefluctuation of a load in such a way as an RCC (ringing choke converter)and the switching frequency fluctuates greatly. It has been verydifficult for the related art to change the on-time of the switchingelement Q2 so as to follow such a fluctuation of the switchingfrequency. In the present preferred embodiment, such a problem is solvedas follows.

In FIG. 13, when a direct-current input voltage, an output voltage, thenumber of turns of a primary winding, and the number of turns of asecondary winding are expressed with Vi, Vo, np, and ns, respectively, arelational expression of ts·Vi/np=tr·Vo/ns is preferably satisfiedbetween a time ts to excite the transformer and a time tr to reset thetransformer, due to the continuity of the magnetic flux of thetransformer. Accordingly, the ratio of the time tr to reset thetransformer to the time ts to excite the transformer may be expressed astr/ts=(ns·Vi)/(np·Vo).

Here, the product, T_(Q1ON)·Ib1, of the discharging time T_(Q1ON) andthe discharge current value Ib1 of the capacitor Cb1 turns out to be adischarging electric charge amount. In the same way, a charging electriccharge amount is expressed as T_(Q2ON)·Ib2. In a steady state, since thedischarging electric charge amount and the charging electric chargeamount are equal to each other, T_(Q1ON)·Ib1=T_(Q2ON)·Ib2 turns out tobe satisfied. Accordingly, the on-time ratioDa=T_(Q2ON)/T_(Q1ON)=Ib1/Ib2=1/Di is satisfied.

Here, when a setting is performed with Di (=1/Da)>(tr/ts), i.e.,Da<(tr/ts), even if the on-time T_(Q1ON) of the switching element Q1 haschanged, the on-time T_(Q2ON) of the switching element Q2 is constantlyshorter than the reset time tr of the transformer. In such a setting, itturns out that the reset time of the transformer is over after theswitching element Q2 has been turned off. In addition to this, theswitching element Q1 is turned on by utilizing the inversion of atransformer voltage due to the finish of the reset of the transformer,and hence, it is possible to cause a current flowing through thetransformer to operate in the current critical mode. Furthermore, evenif the on-time of the switching element Q1 has changed greatly withrespect to the fluctuation of a load, the on-time of the switchingelement Q2 is changed so as to follow this, and it is possible torealize an operation in the current critical mode.

In the present preferred embodiment, a setting is performed so that therelationship of Di=1/Da is preferably satisfied between thecharge-discharge current ratio Di (=Ib2/Ib1) and the on-time ratio Da(=T_(Q2ON)/T_(Q1ON)) and the Di is larger than (ns·Vi)/(np·Vo), and theon-time of the second switching element Q2 is controlled so that theon-time ratio (Da) becomes almost constant with respect to a change in aload current. Accordingly, it is possible to cause the converter tooperate in the current critical mode. In addition, even if the on-timeof the switching element Q1 changes with respect to a change in a loadand a switching period changes, zero-voltage switching is achieved, andit is possible to reduce a loss associated with the reverse recoverytime of the rectifier diode on the secondary side.

By controlling in such a way, it is possible to cause an operationcorresponding to the current-discontinuous mode in the light load stateto be performed in the flyback method in the same way as in the firstpreferred embodiment.

FIG. 14 is a waveform diagram illustrating a relationship among thegate-to-source voltage Vgs1 of the first switching element Q1, thegate-to-source voltage Vgs2 of the second switching element Q2, thedrain-to-source voltage Vds1 of the first switching element Q1, and thevoltage Vcb2 of the capacitor Cb2 in FIG. 13.

When the first switching element Q1 has been put into an on-state, anegative voltage is induced in the high side drive winding nb2, and thecharging voltage VCb2 of the capacitor Cb2 decreases from about 0.6 Vserving as the threshold voltage, for example. After that, when thefirst switching element Q1 has been turned off, a positive voltage isinduced in the high side drive winding nb2, and the charging voltageVCb2 of the capacitor Cb2 increases. When the charging voltage VCb2 ofthis capacitor Cb2 exceeds about 0.6 V serving as the threshold voltage,for example, the transistor Q3 is put into an on-state. Accordingly, thegate electric potential of the second switching element Q2 becomes 0 V,and the second switching element Q2 is turned off.

Since the capacitor Cb2 is charged by the constant current circuit CC21and discharged by the constant current circuit CC22, the slope of theupswing of the charging voltage VCb2 and the slope of the downswingthereof are different from each other. However, even if the on-time ofthe first switching element Q1 has changed, each of the slope of theupswing of the charging voltage VCb2 and the slope of the downswingthereof is constant. Therefore, the on-time ratio Da(=T_(Q1ON)/T_(Q2ON)) is maintained constant due to a preliminarilydefined relationship. In FIG. 14, a waveform diagram when the on-time ofthe switching element Q1 is T_(Q1ON)(1) is indicated by a solid line,and a waveform diagram when the on-time of the switching element Q1changes to T_(Q1ON)(2) is indicated by a dotted line.

Sixth Preferred Embodiment

FIG. 15 is the circuit diagram of a switching power supply apparatus 106according to a sixth preferred embodiment.

The position of the capacitor Cr and the configuration of a secondswitching control circuit 68 are different from the switching powersupply apparatus illustrated in FIG. 13. In this example, a seriescircuit including a resistor R81 and a diode D81 and a series circuitincluding a resistor R82 and a diode D82 are connected in parallel toeach other, and a resistor R7 is connected in series to this parallelcircuit. This series circuit is connected between the high side drivewinding nb2 and the capacitor Cb2. In addition, a circuit where a diodeD11 and a zener diode Dz12 are connected in series to each other in areverse direction is connected between both ends of the capacitor Cb2.

According to this circuit configuration, the output voltage of the highside drive winding nb2 is input, and a constant voltage isbi-directionally generated. In this regard, however, since a chargecurrent with respect to the capacitor Cb2 flows through the resistor R81and a discharge current flows through the resistor R82, a charging timeconstant and a discharging time constant with respect to the capacitorCb2 are different from each other.

In addition, a capacitor C82 is connected in parallel to the diode D82,and hence, this capacitor C82 also causes the charging time constant andthe discharging time constant with respect to the capacitor Cb2 to bedifferent from each other. It is possible to accumulate electric chargein the capacitor C82 during a time period when a backward voltage isapplied to the diode D82, and it is possible to discharge the electriccharge accumulated in the capacitor C82, during a dead time when thevoltage of the high side drive winding nb2 changes. As a result, it ispossible to pass a current whose phase leads compared with a case whereonly a rectifier diode is used. Accordingly, it is possible to adjustthe amount of the current of charge and discharge with respect to thecapacitor Cb2, and, using the capacitor C82, it is possible to correctthe distortion of a charge-discharge current at a dead time, inparticular, when the direction of the charge-discharge current withrespect to the capacitor Cb2 changes.

In addition, the circuit where the diode D11 and the zener diode Dz12,connected between both ends of the capacitor Cb2, are connected inseries to each other in the reverse direction clamps a voltage appliedbetween the base and emitter of the transistor Q3 in a reversedirection, and it is possible to protect an excess voltage from beingapplied.

FIG. 16 is a waveform diagram illustrating a relationship among thegate-to-source voltage Vgs1 of the first switching element Q1, thegate-to-source voltage Vgs2 of the second switching element Q2, thedrain-to-source voltage Vds1 of the first switching element Q1, and thevoltage Vcb2 of the capacitor Cb2 in FIG. 15.

When the first switching element Q1 has been put into an on-state, anegative voltage is induced in the high side drive winding nb2, and thecharging voltage VCb2 of the capacitor Cb2 decreases from about 0.6 Vserving as the threshold voltage, for example. After that, when thefirst switching element Q1 has been turned off, a positive voltage isinduced in the high side drive winding nb2, and the charging voltageVCb2 of the capacitor Cb2 increases. When the charging voltage VCb2 ofthis capacitor Cb2 exceeds about 0.6 V serving as the threshold voltage,for example, the transistor Q3 is put into an on-state. Accordingly, thegate electric potential of the second switching element Q2 becomes 0 V,and the second switching element Q2 is turned off.

While the charging time constant and the discharging time constant withrespect to the capacitor Cb2 are different from each other, a ratiotherebetween is approximately constant. Therefore, the on-time ratio Da(=T_(Q2ON)/T_(Q1ON)) is maintained constant due to a preliminarilydefined relationship.

In addition, in FIG. 15, if necessary, a resistor Rgs to dischargeelectric charge is connected between the gate terminal and the sourceterminal of the second switching element Q2.

Here, in the same way as the description in the fifth preferredembodiment, a case will be described where, after the switching elementQ2 has been turned off, the reset time of a transformer is over, theswitching element Q1 is turned on by utilizing the inversion of atransformer voltage due to the finish of the reset of the transformer,and hence, a current flowing through the transformer is caused tooperate in the current critical mode.

In FIG. 15, when a direct-current input voltage, an output voltage, thenumber of turns of a primary winding, and the number of turns of asecondary winding are expressed with Vi, Vo, np, and ns, respectively, arelational expression of ts·(Vi−np·Vo/ns)=tr·np·Vo/ns is preferablysatisfied between a time ts to excite the transformer and a time tr toreset the transformer, due to the continuity of the magnetic flux of thetransformer. Accordingly, the ratio of the time tr to reset thetransformer to the time ts to excite the transformer may be expressed astr/ts=(ns·Vi)/(np·Vo)−1.

Here, the product, T_(Q1ON)·Ib1, of the discharging time T_(Q1ON) andthe discharge current value Ib1 of the capacitor Cb1 turns out to be adischarging electric charge amount. In the same way, a charging electriccharge amount is expressed as T_(Q2ON)·Ib2. In a steady state, since thedischarging electric charge amount and the charging electric chargeamount are equal to each other, T_(Q1ON)·Ib1=T_(Q2ON)·Ib2 turns out tobe satisfied. Accordingly, the on-time ratioDa=T_(Q2ON)/T_(Q1ON)=Ib1/Ib2=1/Di is preferably satisfied.

Here, when a setting is performed with Di (=1/Da)>(tr/ts), namely,Da<(tr/ts), even if the on-time T_(Q1ON) of the switching element Q1changes, the on-time T_(Q2ON) of the switching element Q2 is constantlyshorter than the reset time tr of the transformer. In such a setting, itturns out that the reset time of the transformer is over after theswitching element Q2 has been turned off. In addition to this, theswitching element Q1 is turned on by utilizing the inversion of atransformer voltage due to the finish of the reset of the transformer,and hence, it is possible to cause a current flowing through thetransformer to operate in the current critical mode. Furthermore, evenif the on-time of the switching element Q1 has changed greatly withrespect to the fluctuation of a load, the on-time of the switchingelement Q2 is changed so as to follow this, and it is possible torealize an operation in the current critical mode.

In the present preferred embodiment, a setting is performed so that therelationship of Di=1/Da is preferably satisfied between thecharge-discharge current ratio Di (=Ib2/Ib1) and the on-time ratio Da(=T_(Q2ON)/T_(Q1ON)) and the Di is preferably larger than(ns·Vi)/(np·Vo)−1, and the on-time of the second switching element Q2 iscontrolled so that the on-time ratio (Da) preferably becomes almostconstant with respect to a change in a load current. Accordingly, it ispossible to cause the converter to operate in the current critical mode.In addition, even if the on-time of the switching element Q1 changeswith respect to a change in a load and a switching period changes,zero-voltage switching is achieved, and it is possible to reduce a lossassociated with the reverse recovery time of the rectifier diode on thesecondary side.

Seventh Preferred Embodiment

FIG. 17 is the circuit diagram of a switching power supply apparatus 107according to a seventh preferred embodiment.

The configuration of a second switching control circuit 69 is differentfrom the switching power supply apparatus illustrated in FIG. 15. Inthis example, a series circuit including the resistor R81 and the diodeD81 and a series circuit including the resistor R82 and the diode D82are connected in parallel to each other, and a circuit where two zenerdiodes Dz1 and Dz2 are connected in series to each other in a reversedirection is connected between the connection point of this parallelcircuit with the resistor R7 and one end of the high side drive windingnb2. That is, the zener diodes Dz1 and Dz2 are added.

In this way, the output of the high side drive winding nb2 is input to aseries circuit including the zener diodes Dz1 and Dz2 and the resistorR7, and a voltage for the series circuit of the zener diodes Dz1 and Dz2is supplied. Therefore, it is possible to significantly reduce andcorrect an influence due to the fluctuation of the voltage of the highside drive winding nb2.

Eighth Preferred Embodiment

FIG. 18 is the circuit diagram of a switching power supply apparatus 108according to an eighth preferred embodiment.

The configuration of a second switching control circuit 70 is differentfrom the switching power supply apparatus illustrated in FIG. 17. Whilea connection relationship between the zener diodes Dz1 and Dz2 andanother connection relationship are different, a function effect isbasically the same as the seventh preferred embodiment.

In this way, the output of the high side drive winding nb2 is input to aseries circuit including the zener diodes Dz1 and Dz2 and resistors R71and R72, and a voltage applied to the zener diode Dz1 or Dz2 is suppliedto a series circuit including the capacitor Cb1 and the resistor R81 orR82. Therefore, it is possible to significantly reduce and correct aninfluence due to the fluctuation of the voltage of the high side drivewinding nb2.

Ninth Preferred Embodiment

FIG. 19 is the circuit diagram of a switching power supply apparatus 109according to a ninth preferred embodiment.

The configuration of a second switching control circuit 71 is differentfrom the switching power supply apparatus illustrated in FIG. 18. In thesecond switching control circuit 71, a capacitor C5 is series-connectedto the resistor R5.

The capacitor C5 and the resistor R5 function as a delay circuit basedon a time constant with relation to applying a voltage to the gate ofthe second switching element Q2, and controls the turnon delay time ofthe second switching element Q2 and a voltage value between the gate andsource thereof, in relation to an input capacitance between the gate andsource of the second switching element Q2.

Tenth Preferred Embodiment

FIG. 20 is the circuit diagram of a switching power supply apparatus 110according to a tenth preferred embodiment.

The position of the capacitor Cr is different from the switching powersupply apparatus illustrated in FIG. 2 in the first preferredembodiment.

Since it is only necessary for the resonance capacitor Cr to be insertedin a path through which a current flowing through the inductor Lr flowsat the time of the off-state of the first switching element Q1, thecapacitor Cr may also be connected between one end of the primarywinding np and the source of the second switching element Q2, asillustrated in FIG. 20.

Eleventh Preferred Embodiment

FIG. 21 is the circuit diagram of a switching power supply apparatus 111according to an eleventh preferred embodiment.

The positions of the capacitor Cr and the inductor Lr are different fromthe switching power supply apparatus illustrated in FIG. 2 in the firstpreferred embodiment.

Since it is only necessary for the resonance capacitor Cr and theinductor Lr to be inserted in series in a path through which a currentflowing through the inductor Lr flows at the time of the off-state ofthe first switching element Q1, the capacitor Cr and the inductor Lr mayalso be connected to both ends of the primary winding np, respectively,as illustrated in FIG. 21.

Twelfth Preferred Embodiment

FIG. 22 is the circuit diagram of a switching power supply apparatus 112according to a twelfth preferred embodiment.

The position of the capacitor Cr is different from the switching powersupply apparatus illustrated in FIG. 2 in the first preferredembodiment.

It is only necessary for the resonance capacitor Cr to be inserted in apath through which a current flowing through the inductor Lr flows atthe time of the off-state of the first switching element Q1. Therefore,the capacitor Cr may also be connected between the drain of the secondswitching element Q2 and the input terminal PI(+), as illustrated inFIG. 22.

Thirteenth Preferred Embodiment

FIG. 23 is the circuit diagram of a switching power supply apparatus 113according to a thirteenth preferred embodiment.

The positions of the second switching element Q2 and the capacitor Crare different from the switching power supply apparatus illustrated inFIG. 2 in the first preferred embodiment.

Since it is only necessary for the resonance capacitor Cr to be insertedin a path through which a current flowing through the inductor Lr flowsat the time of the off-state of the first switching element Q1, thecapacitor Cr may also be connected between the drain of the secondswitching element Q2 and the input terminal PI(G), as illustrated inFIG. 23.

Fourteenth Preferred Embodiment

FIG. 24 is the circuit diagram of a switching power supply apparatus 114according to a fourteenth preferred embodiment.

It is different from the switching power supply apparatus illustrated inFIG. 20 in the tenth preferred embodiment in that capacitors Cr1 and Cr2are provided in addition to the capacitor Cr.

The capacitors Cr and Cr1 are arranged so that the inductor, the primarywinding np, the capacitor Cr, the second switching element Q2, and thecapacitor Cr1 configure a closed-loop.

In addition, the capacitor Cr2 is connected between the connection pointof the capacitor Cr1 with the inductor Lr and the input terminal PI(G).In this way, a plurality of resonance capacitors (Cr1, Cr2) may also beconnected in series to the second switching element Q2.

The capacitor Cr2 is connected, and hence, a current supplied from thepower-supply voltage V1 flows during both time periods of the on-time ofthe first switching element Q1 and the on-time of the second switchingelement Q2. Accordingly, compared with the circuit configuration in FIG.23 where a current flows only during the on-time of the first switchingelement Q1, the effective current of a current supplied from thepower-supply voltage V1 is reduced. Accordingly, it is possible toreduce a conduction loss due to the current supplied from thepower-supply voltage V1.

Fifteenth Preferred Embodiment

FIG. 25 is the circuit diagram of a switching power supply apparatus 115according to a fifteenth preferred embodiment.

The configuration of the secondary side of the transformer T isdifferent from the switching power supply apparatus illustrated in FIG.2 in the first preferred embodiment.

In the fifteenth preferred embodiment, a diode bridge circuit based ondiodes D21, D22, D23, and D34 and a capacitor Co are connected to thesecondary winding ns of the transformer T.

In this way, using the diode bridge circuit, full-wave rectification mayalso be performed.

Sixteenth Preferred Embodiment

FIG. 26 is the circuit diagram of a switching power supply apparatus 116according to a sixteenth preferred embodiment.

The configuration of the secondary side of the transformer T isdifferent from the switching power supply apparatus illustrated in FIG.2 in the first preferred embodiment.

In the sixteenth preferred embodiment, a rectification smoothing circuitbased on a diode Ds and a capacitor Co1 is configured between both endsof the secondary winding ns1 of the transformer T, and a rectificationsmoothing circuit based on a diode Df and a capacitor Co2 is configuredbetween both ends of the secondary winding ns2. In addition, a capacitorCo3 is connected between the output terminals PO(+) and PO(G).

In this way, a voltage doubler rectifier circuit may also be adopted.

Other Preferred Embodiments

While, in each of the preferred embodiments illustrated above, a circuitis preferably provided in which the weight of a load is detected due tothe voltage of the feedback signal, i.e., a current flowing through theload is indirectly detected, a circuit that directly detects a currentflowing through the load may also be provided so as to detect whether ornot being in a light load state. In this case, it is only necessary toadopt a configuration where a control signal is supplied to a switchingcontrol IC so that the blanking function of the switching control IC isactivated at the time of the detection of a light load.

A circuit may also be configured so that the inversion of a windingvoltage is detected due to a voltage change occurring between both endsof the first drive winding nb1 of the transformer T to thus detect theturnoff timing of the second switching element Q2.

In addition, the turnoff timing of the second switching element Q2 mayalso be detected on the basis of a change in the drain-to-source voltageof at least one of the first switching element Q1 and the secondswitching element Q2.

In addition, the turnoff timing of the second switching element Q2 mayalso be detected on the basis of a change in the drain current or thesource current of at least one of the first switching element Q1 and thesecond switching element Q2.

In addition, various preferred embodiments of the present invention maybe applied to a switching power supply apparatus complementarily andalternately turning on and off two switching elements, not only in ahalf-bridge converter but also in a multi-transistor converter such as afull-bridge converter or in a voltage clamp converter or the like, forexample.

In addition, while, in some preferred embodiments illustrated above, anexample has been illustrated where a voltage change occurring betweenboth ends of the winding voltage of the transformer is preferablydetected, a change in the drain-to-source voltage of at least one of thefirst switching element Q1 and the second switching element Q2 may alsobe detected. In addition, a change in the drain current or the sourcecurrent of at least one of the first switching element Q1 and the secondswitching element Q2 may also be detected. In addition, a voltage changeoccurring between both ends of the first drive winding (nb1) may also bedetected.

In addition, the inductor Lr illustrated in some preferred embodimentsillustrated above may also be configured using the leakage inductance ofthe transformer T.

While preferred embodiments of the present invention have been describedabove, it is to be understood that variations and modifications will beapparent to those skilled in the art without departing from the scopeand spirit of the present invention. The scope of the present invention,therefore, is to be determined solely by the following claims.

1. A switching power supply apparatus comprising: a direct-currentpower-supply input unit arranged to receive a direct-current inputvoltage; a transformer including at least a primary winding and asecondary winding magnetically coupled to each other; a first seriescircuit connected to both ends of the direct-current power-supply inputunit and in which the primary winding, an inductor, and a firstswitching element are connected in series; a second series circuitconnected to both ends of the first switching element or both ends of aseries circuit including the primary winding and the inductor and inwhich a second switching element and a capacitor are connected inseries; a switching control circuit configured to put the secondswitching element into an on-state during a predetermined time afterhaving put the first switching element into an on-state during apredetermined time; and a first rectification smoothing circuitconfigured to rectify and smooth an alternating-current voltage outputfrom the secondary winding and output an output voltage to a load;wherein the switching power supply apparatus includes a light loaddetection mechanism configured to directly or indirectly detect a loadcurrent flowing through the load and detect whether or not the load isin a light load state; the switching control circuit includes: a firstswitching control circuit configured to control an on-time of the firstswitching element; and a second switching control circuit configured tocontrol an on-time of the second switching element so that an on-timeratio serving as a ratio of the on-time of the second switching elementto the on-time of the first switching element becomes almost constantwith respect to a change in the load current; wherein in a normal loadstate, an operation is performed in a current-continuous mode where thefirst switching element and the second switching element arecomplementarily and repeatedly put into on-states and off-states; andwhen the light load detection mechanism has detected the light loadstate, an operation is performed in a current-discontinuous mode where acurrent halt time period in which no current flows through thetransformer exists between turnoff of the second switching element andturnon of the first switching element, and a blank time is insertedbetween a time when the second switching element is turned off and atime when the first switching element is turned on next.
 2. A switchingpower supply apparatus comprising: a direct-current power-supply inputunit arranged to receive a direct-current input voltage; a transformerincluding at least a primary winding and a secondary windingmagnetically coupled to each other; a first series circuit in which afirst switching element and a second switching element, connected toboth ends of the direct-current power-supply input unit, are connectedin series; a second series circuit connected to both ends of the secondswitching element and in which the primary winding, an inductor, and acapacitor are connected in series; a switching control circuitconfigured to put the second switching element into an on-state during apredetermined time after having put the first switching element into anon-state during a predetermined time; and a first rectificationsmoothing circuit configured to rectify and smooth analternating-current voltage output from the secondary winding and outputan output voltage to a load; wherein the switching power supplyapparatus includes a light load detection mechanism configured todirectly or indirectly detect a load current flowing through the loadand detect whether or not the load is in a light load state; theswitching control circuit includes: a first switching control circuitconfigured to control an on-time of the first switching element; and asecond switching control circuit configured to control an on-time of thesecond switching element so that an on-time ratio serving as a ratio ofthe on-time of the second switching element to the on-time of the firstswitching element becomes almost constant with respect to a change inthe load current; wherein in a normal load state, an operation isperformed in a current-continuous mode where the first switching elementand the second switching element are complementarily and repeatedly putinto on-states and off-states; and when the light load detectionmechanism has detected the light load state, an operation is performedin a current-discontinuous mode where a current halt time period inwhich no current flows through the transformer exists between turnoff ofthe second switching element and turnon of the first switching element,and a blank time is inserted between a time when the second switchingelement is turned off and a time when the first switching element isturned on next.
 3. The switching power supply apparatus according toclaim 1, wherein the transformer includes a first drive winding, and asecond rectification smoothing circuit that supplies a direct-currentpower-supply voltage to the switching control circuit is provided in thefirst drive winding.
 4. The switching power supply apparatus accordingto claim 1, further comprising: a timing detection mechanism configuredto detect a timing when the second switching element is turned off and awinding voltage of the transformer is inverted; and a feedback circuitconfigured to generate a feedback signal used to detect and control anoutput voltage; wherein the first switching control circuit is a circuitthat causes the first switching element to be turned on with a delayafter the timing detected by the timing detection mechanism and causesthe first switching element to be turned off after a time correspondingto a voltage of the feedback signal has elapsed.
 5. The switching powersupply apparatus according to claim 4, wherein the timing detectionmechanism detects a voltage change occurring between both ends of thewinding voltage of the transformer.
 6. The switching power supplyapparatus according to claim 4, wherein the timing detection mechanismdetects a change in a drain-to-source voltage of at least one of thefirst switching element and the second switching element.
 7. Theswitching power supply apparatus according to claim 4, wherein thetiming detection mechanism detects a change in a drain current or asource current of at least one of the first switching element and thesecond switching element.
 8. The switching power supply apparatusaccording to claim 4, wherein the timing detection mechanism detects avoltage change occurring between both ends of the first drive winding.9. The switching power supply apparatus according to claim 1, whereinthe transformer includes a second drive winding; a connection point ofthe first switching element with the second switching element and afirst end of the second drive winding are connected to each other; thesecond switching control circuit is connected between a second end ofthe second drive winding and a control terminal of the second switchingelement; and the second switching control circuit is a circuit thatcauses the second switching element to be turned on, using a voltage ofthe second drive winding, which occurs due to turnoff of the firstswitching element.
 10. The switching power supply apparatus according toclaim 1, wherein the second switching control circuit is a circuit thatperforms control so that the on-time ratio becomes almost one.
 11. Theswitching power supply apparatus according to claim 9, wherein when theon-time ratio, the direct-current input voltage, the output voltage, thenumber of turns of the primary winding, and the number of turns of thesecondary winding are expressed with Da, Vi, Vo, np, and ns,respectively, the second switching control circuit is a circuit thatcauses the second switching element Q2 to be turned off by performingcontrol so that the Da becomes larger than ns·Vi/np·Vo.
 12. Theswitching power supply apparatus according to claim 9, wherein when theon-time ratio, the direct-current input voltage, the output voltage, thenumber of turns of the primary winding, and the number of turns of thesecondary winding are expressed with Da, Vi, Vo, np, and ns,respectively, the switching power supply apparatus is a circuit thatcauses the second switching element Q2 to be turned off by performingcontrol so that the Da becomes larger than ns·Vi/np·Vo−1.
 13. Theswitching power supply apparatus according to claim 1, wherein the lightload detection mechanism performs detection using a voltage of thefeedback signal.
 14. The switching power supply apparatus according toclaim 1, wherein the light load detection mechanism performs detectionbased on whether or not a switching frequency for driving the firstswitching element or the second switching element has reached an upperlimit value.
 15. The switching power supply apparatus according to claim1, wherein in the current-discontinuous mode, along with a decrease inthe load current, the switching control circuit causes a switchingfrequency to drive the first switching element or the second switchingelement to decrease.
 16. The switching power supply apparatus accordingto claim 1, wherein in the current-discontinuous mode, along with adecrease in the load current, the switching control circuit reduces theon-time of the first switching element or the second switching element.17. The switching power supply apparatus according to claim 1, whereinin the current-continuous mode, the switching control circuit sets amaximum level of the switching frequency or a maximum value of theon-time of the first switching element or the second switching element.18. The switching power supply apparatus according to claim 1, whereinin the current-discontinuous mode, the switching control circuitcontrols the switching frequency or the on-time of the first switchingelement or the second switching element, in response to the feedbacksignal.
 19. The switching power supply apparatus according to claim 1,wherein the switching control circuit sets a limit value on a voltage ofthe feedback signal, and causes switching of the first switching elementor the second switching element to be halted when the voltage of thefeedback signal has exceeded the limit value.
 20. The switching powersupply apparatus according to claim 19, wherein the switching controlcircuit operates with a lowest switching frequency and a minimumon-time, with respect to the limit value.
 21. The switching power supplyapparatus according to claim 1, wherein in the current-discontinuousmode, while the switching control circuit does not cause the secondswitching element to be turned on even if the first switching elementhas been turned off, the switching control circuit applies a forwardcurrent to a parasitic diode of the second switching element first, andapplies a backward current during a subsequent reverse recovery time.22. The switching power supply apparatus according to claim 1, whereinat least one rectifying element in the first switching element, thesecond switching element, or the first rectification smoothing circuitis a field-effect transistor.
 23. The switching power supply apparatusaccording to claim 1, wherein the inductor includes a leakage inductanceof the transformer.